The forward converter shown in FIG. 2a transfers the power through the isolation transformer to the load during the interval DTS when the primary side switch S1 is ON. The prior-art converter of FIG. 2b transfers the power during the complementary interval D′TS when switch S′2 is ON. Note that the primary side switching is identical in both converters. The DC conversion gain of each converter is also same and equal to DVg—where D is duty ratio of the main switch, except for the inverting polarity of the transformer secondary for the converter of FIG. 2b. Hence, another prior-art, two-output converter can be constructed, which has two secondaries, as in FIG. 6. One DC output is as in converter of FIG. 2a and another DC output has identical DC voltage as converter of FIG. 2b. Because of the common DC voltage gain, the two-output converter of FIG. 6 can be combined into a single output converter of FIG. 7 with substantial circuit simplification: the two secondary side transformer windings can be merged into just one secondary winding, which, in turn, makes possible merging of two S2 rectifier switches into a single one, since the energy transfer interval for S2 switch and inductance L1 of first output is at the same time freewheeling interval for using the same S2 switch for inductance L2. The same is true for the other switch S′2 thus resulting in a prior-art converter of FIG. 7 with only two switches, but now with two output inductors, which is often referred to as a Current-doubler converter.
If the two outputs of converter in FIG. 6 are connected, the DC load current will be split between two secondary sides based on two resistances R1 and R2. From the above derivation, one of the key characteristics of this converter is obtained: the DC load current of the converter of FIG. 7 will be split according to the DC impedances at the two output inductors, that is R1 and R2 resistances. In a special case when R1=R2 the DC load current is split equally between two outputs and I1=I2=0.5I. Therefore, the more appropriate name for the converter of FIG. 7 would be current divider as it divides the well-known quantity DC load current I into half, so that I/2 is flowing through each of the two separate inductors.
The converter of FIG. 7 has negative output voltage polarity if the common rectifier connection (cathode connection) is connected to a secondary ground. Shown in FIG. 8a is a positive polarity alternative in which common rectifier connection is the anode connection of two diodes. This is preferred implementation for positive output voltage, as the synchronous rectifier MOSFET transistor replacing rectifier diodes, will be in a preferred grounded source connection relative to output ground. However, unlike other prior-art converters, the converters in FIG. 7 and FIG. 8a have three magnetic components. Worse yet, each of the three magnetic components has a substantial DC energy storage and requires a large air-gap to sustain DC load current bias as illustrated in FIG. 8b for the two output filtering inductors. This results in more complex filter than in forward converter, as there are now two inductors instead of one.
DC Analysis
The following DC analysis of the converter in FIG. 8a applies equally well to the present invention due to their equal DC characteristic.
The converter operates at the fixed switching frequency given by fS=1/TS where TS is the switching period. Two of the four switches are ON for duty ratio D defined as D=tON/TS, and two other switches are ON during remaining time of the switching period TS, defined as D′=1-D.
By use of the state-space averaging method (S. Cuk, “Modelling, Analysis, and Design of Switching Converters, Ph.D. thesis, California Institute of Technology, November 1976; R. D. Middlebrook and S. Cuk, “Advances in Switched-Mode Power Conversion”, Vol. I, II, and III, TESLAco), and assuming 1:1 transformer turns ratio, the output DC voltage is obtained asV=DVg  (1)
As is well known to those skilled in the art, the turns ratio of the transformer can then be used to obtain additional step-down of the input voltage. Hence, this converter belongs to the class of the fixed switching frequency Pulse Width Modulated (PWM) switching converters with the buck-type step-down DC conversion ratio characteristic.
First we analyze how is the DC load current I divided between the two inductors, L1 with DC current I1 and L2 with DC load current I2. From the DC currents summation at the node A we haveI=I1+I2  (2)where the split of DC load current I into two components depends on the second order parasitic effects that is the respective DC resistances R1 of the inductor L1 and R2 of inductor L2. In case R1=R2 the DC load current is equally divided between two inductorsI1=I2=I/2  (3)
Let us now analyze the load current I in general case of arbitrary split, that is for I1 and I2 given by (2). In that case the following relationship must be satisfied:R1I1=R2I2  (4)
When switch S2 is open, and S′2 closed, the current at node A splits into two parts so that during D′TS interval DC current I1 flows through inductor L1, while DC current I2 flows through and into dotted terminal of secondary winding of the transformer.
When switch S′2 is open, and S2 closed, the current at node B splits into two parts so that during DTS interval DC current I2 flows through inductor L2, while DC current I1 flows through and out of the dotted terminal of single turn secondary winding of the transformer.
The above analysis of DC load current distribution applies equally well to all Integrated Magnetics variants of the present invention as the magnetics integration does not alter DC gain properties of the converter.
Output Ripple Current Characteristic
With the designation of the positive ripple current directions and respective AC voltages on two inductors v1 and v2 as in FIG. 8a ripple currents Δi1 and Δi2 in two inductors and Δi in the output can be evaluated asΔi1=TSv1/L1  (5)Δi2=TSv2/L2  (6)Δi=Δi1−Δi2=TS(v1/L1−v2/L2)  (7)since the output ripple current is the difference of the two inductor ripple currents.
As seen from general AC analyses in later section for a special case of 50% duty ratio, v1=v2=0.5 v. Therefore, the output ripple current from (7) becomes zero. As graphically displayed in FIG. 8c the two inductor current ripples, however, are still very large and may be even 50% or more of the DC load current. Therefore, zero ripple current (at 50% duty ratio) and a proportionally increasing output ripple current away from 50% duty ratio are obtained as a difference of two large AC ripple currents.
This zero ripple feature at one nominal point is a distinct advantage over the Current-doubler converter. However, the disadvantage is that two separate large magnetic cores are needed for inductances as seen in FIG. 8b. 